Topology Selection By the Numbers Part Two

Apr 1, 2006 12:00 PM
By Ernie Wittenbreder, Chief Engineer, Technical Witts, Flagstaff, Ariz.

In this second part of a three-part series, we consider some of the alternatives to the bucks and boosts discussed in part one for nonisolated stepup and stepdown conversion for situations in which there may be better choices.

For a PDF verion of this article, click here.

In part one of this article series, we found that total component stress factors (CSFs) — a new analysis method that enables the comparison of different power conversion topologies — for buck and boost converters were the same for fixed-line voltage where the stepdown ratio of the buck converter was equal to the stepup ratio of the boost converter. In this second part, we focus on nonisolated applications with limited line-voltage range.

Buck and boost converters are part of the same canonical cell where the differences between the buck circuit and the boost circuit are the line and load connections. We illustrate this relationship in Fig. 1 where the circuit operates as a buck converter for power flow from left to right and as a boost converter for power flow from right to left.

Instead of treating the stepup and stepdown problems separately, we recognize that a stepup converter is a stepdown converter that operates with power flow reversed. Therefore, any of the circuits in this second part of the article series can be operated either as stepup or stepdown. In every case, power flow from left to right will correspond to stepdown conversion, and power flow from right to left will correspond to stepup conversion.

In part one, we found that the lowest CSFs for the buck converter occurred for stepdown ratios near 1 and that the lowest CSFs for the boost converter occurred for stepup ratios near 1. We will consider the problem where the stepup or stepdown ratio is near but not equal to 1, as well as some alternatives to simple bucks and boosts.

Fig. 2 illustrates a tapped-inductor buck or boost, which is an attractive candidate for small stepup and stepdown ratios. The only difference between this converter and the simple buck and boost of Fig. 1 is the second winding of the choke placed in series with the lower MOSFET (M_LOWER). The circuit in Fig. 1 is contained in the Fig. 2 circuit and represents the case in which the turns ratio of L_LOWER to L_UPPER is zero.

For Fig. 2, there will be a different set of CSFs for each turns ratio, and the turns ratio can be varied to achieve different results. We will consider the case where the load/line at the right side of the circuit is 380 V and the line/load at the left side of the circuit is 400 V at a power level of 400 W. We also will consider the minimum-voltage tapped-inductor converter^[1], shown in Fig. 3, for the small ratio-conversion problem.

The Fig. 3 circuit is of minimum voltage in the sense that none of the transistors has voltage stress higher than the load/line voltages. Refer to part one of this article series for information on how to calculate the CSF for each component. Table 1 summarizes optimized results for Figs. 1, 2 and 3.

For the Fig. 2 circuit, if we optimize component weights for minimum total semiconductor component stress factor (SCSF), we find that the optimal turns ratio is 17.9 and we achieve a 20 times reduction in total semiconductor stress but at a cost. With a 17.9 turns ratio, the total winding component stress factor (WCSF) has almost doubled and the capacitor component stress factor (CCSF) has increased by almost two orders of magnitude.

In many applications, this may be a good tradeoff since the input capacitor requirement may be driven by an electromagnetic interference filter requirement or an interface stability issue, such as Middlebrooke's rule, and the output capacitor requirement may be driven by a load transient or ripple voltage requirement.

Table 2 also illustrates a compromise case where the turns ratio is selected to be 1-to-1. With a 1-to-1 turns ratio, a substantial improvement in total SCSF is realized with more modest degradations in total WCSF and total CCSF. The lower semiconductor component stresses are achieved by using a low-voltage MOSFET for M_UPPER.

With a 17.9 turns ratio, M_UPPER has a voltage stress of just slightly greater than 40 V. The voltage stress on M_UPPER can be large during a startup transition in the buck implementation and during a shutdown transition in the boost implementation, without some mechanism to protect M_UPPER while the load/line voltage to line/load voltage differential is high.

Table 1. Optimal CSF values for small step-ratio converters.

	Figs. 1 and 2 Turns ratio = 0	Fig. 2 Turns ratio = 17.9	Fig. 2 Turns ratio = 1	Fig. 3 Turns ratio = 11.1
MUPPER SCSF	1.29	0.04	0.4	0.036
MLOWER SCSF	0.3	0.038	0.12	0.019
MRESET SCSF	0	0	0	0.021
Total SCSF	1.59	0.078	0.52	0.076
LUPPER WCSF	0	0.01	0.01	0.007
LLOWER WCSF	0.01	0.009	0.002	0.005
Total WCSF	0.01	0.019	0.012	0.012
CIN/OUT CCSF	0.053	1.89	0.16	3.01
COUT/IN CCSF	0	1.7	0.07	2.76
CMIDDLE CCSF	0	0	0	0.15
Total CCSF	0.053	3.59	0.23	5.92

Table 2. Optimal CSF values for large step-ratio converters.

	Figs. 1 and 5 Turns ratio=0	Fig. 4 Turns ratio=5	Fig. 5 Turns ratio=9.14	Fig. 5 Turns ratio=3	Fig. 6 Turns ratio=5
MUPPER SCSF	40	1.61	13.4	13.7	6.4
MLOWER SCSF	120	2.96	12.6	20.6	11.8
MRESET SCSF	0	1.64	0	0	6.5
Total SCSF	160	24.8	26.1	34.3	24.8
LUPPER WCSF	0	0.4	2.9	1.7	1.6
LLOWER WCSF	3.24	0.56	3	3.4	2.24
Total WCSF	3.24	3.84	5.9	5.1	3.84
CIN/OUT CCSF	9	0.39	1.7	2.9	2.3
COUT/IN CCSF	0	0.48	1.5	0.9	2.9
CMIDDLE CCSF	0	0.48	0	0	2.9
Total CCSF	9	1.35	3.3	3.8	8.1

One way to solve this problem is by keeping M_UPPER in its conducting state during the high differential voltage transition. All of the tapped-inductor power converters in this article achieve efficiency improvements by using MOSFETs with lower voltage ratings than the peak dc voltages present in the circuit. One must consider the potential for the application of voltages higher than the steady-state voltages to low-voltage MOSFETs during voltage transients, and implement necessary avoidance or protection mechanisms. Often, a single zener diode can eliminate the problem.

For a topology that has superior total SCSF and superior total WCSF, a high CCSF can be dealt with by employing an interleaved multiphase configuration. Paralleling multi-phase subconverters has no effect on total SCSF or total WCSF, but multiphasing reduces total CCSF by a factor equal to or greater than the square of the number of parallel subconverters (Fig. 4).

In the Fig. 3 circuit, M_RESET operates in synchronization with M_UPPER. C_MIDDLE is charged during the time that M_LOWER conducts and is discharged when M_UPPER and M_RESET are on. Results for the Fig. 3 circuit indicate that it provides SCSF improvements that are slightly better than those achieved by the Fig. 2 circuit, but with much lower total WCSF and an even greater increase in total CCSF. There are practical zero-voltage switching (ZVS) variations for each of the figures illustrated, but the Fig. 3 circuit requires no additional active switches to accomplish ZVS. Another topology that might be a candidate for small step ratios is the isolated flyback converter with the secondary circuit connected to line and load but not to ground.

Another problem where candidate topologies exist that compete well with simple bucks and boosts is the problem of large step ratios. We consider the problem of stepping from 100 V to 1000 V and from 1000 V to 100 V in a 100-W converter. We will not consider the effects of wide line-voltage variations at this time. The candidate topologies that we will consider are the buck and boost of Fig. 1, the tapped-inductor buck and boost for large step changes (Fig. 5), the minimum-voltage tapped-inductor buck and boost for large step changes (Fig. 6) and a multiphase implementation of the Fig. 6 circuit in which two parallel subconverters share the same set of three capacitors, illustrated in Fig. 4.

Table 2 illustrates the results achieved. The simple buck and boost converters of Fig. 1 have high SCSFs because the switches have both high voltage stress and high current stress. In the Fig. 5 circuit optimized for minimum total SCSF, M_UPPER has high voltage stress (slightly less than 2 kV) but low current stress, and M_LOWER has high current stress but low voltage stress (slightly less than 200 V). The circuit in Fig. 5 achieves a 6-to-1 reduction in total SCSF compared to that in Fig. 1. The Fig. 5 circuit also achieves a reduction in total CCSF, but total WCSF increases by almost a factor of 2.

The Fig. 6 circuit achieves slightly lower total SCSF than the Fig. 5 circuit and significantly lower WCSF, only about 20% higher than the simple buck and boost, but the total CCSF is only slightly lower than the Fig. 1 circuit and significantly higher than the tapped-inductor circuit of Fig. 5. In the Fig. 6 circuit, the voltage stresses of M_UPPER, M_LOWER and M_RESET are 1000 V, 167 V and 833 V, respectively. We address the issue of high CCSF by implementing a two-phase converter based on the Fig. 6 circuit in which the capacitors are shared between the two subconverters.

We use the rules presented in part one of this article series to calculate the total SCSF and total WCSF for the dual subconverter combination. Since we only have one set of capacitors, we must calculate CCSFs for each capacitor again, considering that the capacitor currents are a combination of currents from the two subconverters. Fortunately, the capacitor currents from the two subconverters cancel each other to a large extent due to the fact that the subconverters are operated 180° out of phase. For the MOSFET M_UPPER1 the peak voltage stress is 1000 V, the on-state current is 0.0835 A and the duty cycle is 0.599. The root-mean-square (RMS) current in M_UPPER1 is:

By optimizing for minimum total SCSF, we find that the optimal weight for M_UPPER1 is 0.992 and the sum of the semiconductor weights is 3.822. The SCSF for M_UPPER1 is:

For M_LOWER1 the SCSF is 2.96 and for M_RESET1 the SCSF is 1.64, so that the total subconverter SCSF is 6.2. Since the second subconverter is identical to the first except for operating phase, its total subconverter SCSF is also 6.2. For the total SCSF for a combination of subconverters in part one of this article series, we defined:

where W_I is the semiconductor subconverter weight for subconverter I, is the sum of semiconductor weights and Total SCSF_I is the total SCSF for subconverter I. We want the two subconverters in a multiphase arrangement to be equal in every respect except phasing, so we will assign a semiconductor subconverter weight of 1 to each subconverter.

In this case:

For the combination of subconverters the total SCSF is:

which is the same result we had for Fig. 6. We would expect there to be no advantage or disadvantage to breaking the converter into two parallel subconverters — each operating at half power — and the calculated total SCSF confirms that fact.

For L_LOWER2 the applied voltage is:

The RMS current in L_LOWER2 is given by:

The weight assigned to L_LOWER2 is 1.4 and the sum of the winding weights is 2.4. The CSF for L_LOWER2 is:

For L_UPPER2 the WCSF is 0.40 and the total WCSF for the subconverter is 0.96. We assign a winding subconverter weight of 1 to each subconverter so that the sum of the winding subconverter weights is 2. For the combined system:

which is identical to the total WCSF obtained for the single-phase circuit, as expected.

Finding the total CCSF for the Fig. 4 circuit presents a different problem, because the three capacitors shown in Fig. 4 are shared by two subconverters. We calculate each CCSF considering that the capacitor currents are formed by algebraic sums of currents from the two subconverters. We assume that subconverter 1 is beginning its operating cycle and subconverter 2 is halfway through its operating cycle.

The average line/load current is P/V_LINE/LOAD, or 0.1 A. For C_IN/OUT the capacitor current is the difference between the average line/load current and the sum of currents in M_UPPER1 and M_UPPER2. At the beginning of an operating cycle for a duty cycle of (0.599 - 0.500) = 0.099, both M_UPPER1 and M_UPPER2 conduct so that the C_IN/OUT net current is 0.0835 A 1 0.0835 A 2 0.1 A = 0.067 A. In the subsequent phase of the operating cycle, M_UPPER2 turns off and the net current is 0.0835 A 2 0.1 A = - 0.0165 A. The duty cycle for the second phase is 0.401.

During the next phase of the operating cycle, with duty cycle equal to 0.099, M_UPPER2 is on again and the net C_IN/OUT current is 0.067 A. During the last phase, M_UPPER1 is off, the duty cycle is 0.401 and the current is -0.0165 A. The RMS current in C_IN/OUT is 0.033 A. To achieve optimal (minimal) total CCSF, a weight of 0.964 is assigned to C_IN/OUT. The sum of capacitor component weights is 3.364. We can now calculate a CCSF for C_IN/OUT:

For C_OUT/IN and C_MIDDLE the CCSF is 0.48, and the total CCSF for Fig. 4 is 1.346. Among the many benefits of parallel multiphase converters is much-reduced capacitor component stress, which translates directly into higher efficiency. Compared to Fig. 6, Fig. 4 has improved total CCSF by a factor of 6. In general, multiphasing improves total CCSF by a factor equal to or greater than the square of the number of subconverters, if the topology enables capacitor sharing. Comparing Fig. 4 with Fig. 1, we achieve much-improved total SCSF and total CCSF, for a 20% increase in total WCSF, but with a much larger component count.

With sets of operating equations for candidate circuit topologies and a spreadsheet or other mathematical computation program, one can readily generate a set of numbers for numerical topological comparison using the CSF method. An Excel spreadsheet illustrating the calculations in this article is available at www.TechnicalWitts.com. Next month, in the third and final part of this article series, we will compare isolated topologies and consider the effects of wide line-voltage range on power converters.

References

1. Wittenbreder, E.H., “Tapped Inductor Power Conversion Networks,” U.S. Patent Application 60/757561. Available online at www.technicalwitts.com.

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