Sensorless Motor Control Simplifies Washer Drives

Jun 1, 2006 12:00 PM
By Aengus Murray, Director iMOTION Products, Energy Savings Product Group, International Rectifier,

Sensorless Control

A sensorless control algorithm can use motor-winding-current measurements to indirectly determine the rotor-flux position. This algorithm allows for sinusoidal currents to drive the motor. These sinusoidal waveforms produce low torque ripple, which minimizes the acoustic noise. The control method also easily provides for field-weakening operation for high-speed spinning.

Typically, sensorless control algorithms determine the rotor-flux position based on the winding's back EMF — an indirect measurement that derives from the motor circuit model. The advantage of this approach is that it is applicable when sinusoidal-current waveforms drive the motor, which produce smooth, glitch-free torque.

The control diagram in Fig. 2 describes the field-oriented-control (FOC) algorithm for the permanent magnet synchronous motor (PMSM). A feature of this algorithm is that it transforms the three-phase ac currents in the stator windings into a rotating reference frame where they appear as two dc-current components representing the stator current's torque and flux components.

The reference transformations occur in two stages. The first stage calculates the equivalent stator currents in a two-phase machine model. The second performs a vector rotation to calculate the stator currents' effect in a reference frame synchronous with the rotor shaft.

The two-phase stator-current model forms the basis for the rotor-flux-estimation algorithm, which is defined as:

where ψ_R is the rotor flux.

As shown in Fig. 3, the inputs are the stator voltages (v_α and v_β) that the drive system provides and the stator-current measurements (i_α and i_β). Calculations for the winding voltage's back-EMF components use the stator-current measurements as input. Integration of the back-EMF components produces two rotor-flux components that are sine and cosine functions of the rotor angle.

The second stage in the rotor-angle estimator is a phase-locked loop that forces the sine and cosine of the angle estimate to track the sine and cosine rotor-flux functions, respectively. This approach has the advantage of delivering both rotor-angle and velocity estimates, and is independent of the rotor-flux magnitude.

The controller performs the current-control loop calculations within the rotating reference frame and compensates for the winding resistance, inductance and back-EMF magnitude. The input to the torque current loop comes from the outer velocity-control loop that compensates for the mechanical load's dynamics. The field-weakening controller sets the flux-current loop input to zero at low speeds to maximize the output torque.

At the motor's base speed, the back EMF due to the PM flux reaches the dc bus voltage. At this speed, drives without field weakening would be unable to further increase the motor speed because, with the back EMF equal to the dc bus voltage, there is no excess voltage available with which to drive additional current.

However, the field-weakening controller can provide a negative input to buck a fraction of the PM flux that couples to the winding. This allows the field-weakening controller to set motor speeds beyond the base speed. The advantage of the sensorless FOC implementation is that it delivers smooth torque with good dynamic control — without the use of any position sensors.

Algorithm Implementation

The hardware platform for the washing-machine controller consists of a digital control IC and an integrated power module, as appears in the application circuit in Fig. 4. An embedded 16-bit motion-control engine (MCE) implements the sensorless control algorithm that performs all of the control calculations using hardware macro blocks.

There is an independent 8-bit microcontroller for the application software that communicates with the MCE using shared memory. The MCE connects directly to the motor-control peripherals that interface to the inverter power module that drives the motor.

The PWM unit calculates the duty-cycle timing of the power transistors to control the voltage applied to each phase of the motor while the analog-to-digital converter (ADC) samples the motor currents flowing in the dc-link shunt resistor. The high-voltage IC (HVIC), gate-drive circuit interfaces between the 3.3-V logic-level outputs of the digital control IC and the power transistors. The HVIC also includes inverter-protection features such as overcurrent shutdown that also use feedback information from the dc-link shunt.

The digital implementation of the FOC algorithm that appears in Fig. 5 has the same basic structure as that in Fig. 2, with an outer velocity loop and two inner current loops for controlling the motor torque and flux. The other control functions include the rotor-angle estimation, field weakening for high-speed operation and the phase-current reconstruction block. Practical implementations need additional functions such as those that the startup-sequencer and fault-detection blocks provide.

Each control-loop compensator has a proportional-plus-integral (PI) block with a limiting function at its output. The output of the velocity loop, which is the reference input to the torque loop, is limited so that the controller does not drive too much current into the motor. The controller limits the current-loop outputs to a voltage range that the power inverter can deliver. An output from each limit block halts the integration upon reaching the limit, to prevent unwanted integrator windup while the controller is saturated.

The vector-rotation block (e^jθ) transforms the current-loop outputs to ac quantities in the stator-reference frame. The two-phase stator-reference voltages directly feed the space-vector PWM block that controls the voltages to the motor since the modulation algorithm has an embedded two- to three-phase transformation.

The most difficult part of the rotor-angle-estimator implementation is in the low-speed range. The flux-estimator algorithm's integrators have a low frequency cutoff to avoid saturation due to dc offsets. At start up, the back EMF is zero and so the motor must start without position feedback.

The first stage of the startup algorithm parks the rotor in known position by driving dc currents into the stator windings. In the second stage, the rotor flux estimator is in open-loop mode, and the estimator's angular velocity output signal increases at a constant rate. In this mode the torque-reference current is a constant value to deliver a torque that will accelerate the motor at the rate equal to the angular velocity output of the rotor-flux estimator.

If the motor torque is greater than that which the system requires to match the estimator acceleration, the rotor will advance in phase and move out of optimum alignment. However, since the motor torque function is a cosine function of this alignment, the rotor angle advances until the motor torque exactly matches the target value.

At 5% of the rated velocity, by which point a measurable level of back EMF develops, the estimator switches into the closed-loop mode. The feedback loops are then controlled by the estimated angle and estimated speed signals from the rotor-flux estimator.