A prototype step-down resonant converter employs zero-voltage and zero-current switching while offering a 4-kW output with a 280- to 440-Vdc input and 14-Vdc output.
Depending on the type of regulation they use, step-down dc-dc converters usually fall into two groups: fixed frequency with pulse-width modulation (PWM) or variable frequency. Converters with PWM provide good regulation from minimum power (<5% of nominal power) to maximum power with good efficiency, but they have disadvantages. They are difficult to control at small pulse widths, have relatively high power consumption under idle or light-load conditions (below 5% of nominal power), and also generate EMI. Solving these technical problems usually leads to cost increases.
Resonant topologies with variable frequency generate relatively little EMI, can run at no load (depending upon topology), have low power consumption at idle (again, depending upon topology), and their cost is reasonable. However, it is also more difficult with a resonant converter to achieve an input voltage range of more than: Vmax/Vmin =1.3.
At the present time, topologies using PWM by phase-shift control are popular for basic high-power applications of 4 kW or more. These topologies reduce EMI because they use soft-switching technology. Small pulse widths can also be generated, but hard switching occurs at light load (unless complexity is added), increasing losses and EMI. More problems occur when requirements call for low output voltages (14 Vdc) and high currents (200 Adc and higher).
In this case, using the power inductor in the rectifier circuit (Refs. 1, 2, 3, 4, 5) will be a problem for three reasons:
- The power losses in the inductor are large.
- It will be difficult to provide soft-switch for the rectifier diodes.
- The cost of the inductor significantly impacts BOM.
Several designs (Refs. 6, 7, 8, 9, 10, 11) have gone a long way toward partially solving these issues. Some have a greater number of active switches, and in some the commutation is not purely zero-voltage switching (ZVS). In addition, it is not clear how they would behave under idle or light-load conditions. Finally, there is a question as to whether they would still achieve good efficiency with power levels of 2 kW or more.
The circuit described here uses the advantages of both PWM and variable-frequency topologies to overcome these difficulties and is relatively cost effective. This topology uses PWM and fixed (maximum) frequency of commutation (up to approximately 200 kHz) when output power is between 25% and 100% of rated. When output power is less than 25% control in the converter is accomplished by reducing the commutation frequency and continuing to reduce the PWM duty cycle. The commutation frequency can reach 2 to 3 kHz and the PWM is 0% duty cycle at idle, hence achieving very low idle losses (maximum 8 W). This is very important for battery-powered applications.
Fig.1 shows this converter's topology, which is bi-directional, but we will only discuss step-down operation. Refs. 12 and 13 describe how this topology works in step-up mode. The converter has a series resonant topology so the conversion process delivers energy packets of limited size. The energy-packet size is controlled by the resonant tank of L RES and capacitors C1 through C4, as well as the conversion pulse width. The maximum commutation frequency in this topology is:
Fig.2 shows theoretical waveforms of the converter when the commutation frequency is at maximum and PWM provides the regulation. Figs. 3a to 3f show equivalent circuits of each operation mode. Start with Fig. 3a, which shows the time segment t0 to t1. At t0 the devices S2, D3, and S4 have just begun conduction. They started conduction with zero current and the current in the circuit rises sinusoidally. The value and waveform of this current is determined by the resonant circuit (C1 to C4 and L RES) and the value of load.
At t1, switch S4 turns off and interrupts the current while, at the same time, S3 turns on and prepares for the next half-cycle. This interruption happens under ZVS conditions because S3 and S4 are located between capacitors C1 and C2 on one side and C3 and C4 on the other side. C3 and C4 have significantly smaller values than C1 and C2. The purpose of C3 and C4 is to provide soft-switch commutation for S3 and S4 between t1 and t2 (Fig.3b).
At t2 the energy-discharge process of inductor LRES begins. Between t2 and t3 current flows in the circuit D6, LRES, Tr and S2 (Fig.3c). This current reduces linearly to zero in time:
n = Turns ratio of the transformer.
I = Peak current at t2
At t3 the current flow in the low-voltage side in D7 and D10 has decayed to zero. Between t3 and t4 on the high-voltage side, the magnetizing current flows via circuit S2, TR, LRES and D6 (Fig.3d). At t4 switch S2 turns-off, interrupting the magnetizing current, which has a relatively small value. Therefore, the body capacitance of S2 provides ZVS turn-off. Between t4 and t5 (Fig. 3e) magnetizing current charges the body capacitance of S1 and S2 and prepares S1 for ZVS turn-on.
At t5, S1 turns on and a new half-cycle of power conversion begins, which is identical to the half-cycle between t0 and t5. Between t5 and t6 current flows via S1, Tr, Lr, S3, D4 and C1-C4 (Fig.3f).
How to calculate the values of the resonant components can be found in (Ref. 12), where the authors of this paper described the step-up operational mode of this topology. To achieve good performance, the Q of the resonant inductor should be greater than 7.
A prototype was built for an output power of 4 kW, an input voltage range of 280 Vdc to 440 Vdc and an output voltage of 14 Vdc. Fig.4 shows the basic schematic.
Q1, Q2 are IXSN80N60BD1
Q3, Q4 are IXKN75N60C
Q5, Q6, Q7, Q8 are IRF2804, 7 in parallel
D1, D2 are DSEI2×101-06A
MOSFETs (Q5 to Q8) make up a synchronous rectification.
Fig.5 and Fig.6 show actual waveforms of the current and voltage for the 4-kW prototype at a 400-Vdc input and a 14.2-Vdc output with a 280-Adc load. It is clear that the theoretical predictions of the waveforms are very close to the practical results.
Fig.7 shows the waveforms when the converter produces 540 W of output. We see how the commutation frequency has started to change; the current scale is 10 A/div. Fig.8 shows measured data taken from the 4-kW prototype under different operating conditions. The converter's efficiency is above 80% under light load (200-W output, 5% of nominal power).
The Table summarizes a comparison between the new topology and two topologies that use phase-shift control. Fig. 9 shows the standard phase-shift-controlled topology, and Fig. 10 shows an improved variant of that topology [Ref. 1]. We did this comparison with one assumption: all topologies have the same efficiency, output power, and input and output voltage.
Let us examine these comparisons. When the new topology produces nominal output power, the voltage that appears across the winding of the transformer has approximately a trapezoidal waveform with roughly constant RMS voltage regardless of input voltage. The current carried by the winding is almost sinusoidal with a duty cycle of 90% to 95% and does not change with input voltage. This is an advantage when considering transformer losses and, along with the fact that the new topology can reach higher commutation frequency, significantly helps to reduce transformer size, increase transformer efficiency and decrease price. The benefits in operating frequency are driven by the lack of rectifier recovery; since the new topology is resonant, rectifier recovery time does not contribute to circuit loss.
The phase-shift topology runs into hard switching at light load unless additional circulating energy is provided to commutate the switches. The new topology has much less capacitance to commutate at light load. The benefits of continuous soft switching of all of the semiconductors and lower transformer losses will make this topology a more reliable configuration.
When the new topology operates under PWM control it works like a current source because, during this time, the resonant inductor is discharging energy into the load via the transformer (time period t1 to t3, Fig.2). This high-impedance condition and strong control through energy packets help when we parallel two or more of these power stages, leading to good sharing of current without any special current-sharing control needed. The impedance is also helpful when using the new topology as a power-factor corrector with an isolation transformer.
It is important to note that the simplicity of the rectifier in the new topology significantly contributes to cost reduction, therefore leading to greater cost effectiveness. Although the control for this new topology is more complex it should be pointed out that if the control used in this topology is implemented in an integrated fashion, such as with an ASIC, the cost of the control will decrease and its use will be much less of a factor in overall cost than the benefits accrued in the power stage.
The new topology has a minimum of 10% lower cost than the standard phase-shift control topology at the present time. When the cost of the control reduces, this number will increase. In other words, the cost of the topology will go down even more. One further benefit of this new topology is that it can readily be adapted to bi-directional operation, as can most other topologies; however in this case both directions can be operated with soft switching (Refs. 12, 13). This allows for the design of a bi-directional converter that is not hampered by the operating conditions in any particular conversion direction. In other words up-conversion will have a similar efficiency to down conversion.
The topology presented here has some notable advantages compared with a typical phase-shift-controlled topology, notably: it uses all of the power components efficiently, all of the switches have low switching losses, and can work with any rectifier topology without an inductor. Power consumption is very low under idle or light-load conditions, which makes this topology very suitable for battery-powered applications. Bi-directional operation is readily achieved with high performance in both directions, offering particular benefits in specialist applications. Its wide input-voltage range also makes this topology a good candidate for applications such as power-factor correction. The new topology also suits a dc-dc converter at input voltages greater than 200 Vdc and power levels greater than 2 kW.
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